Current control circuitry

ABSTRACT

The present disclosure relates to current control circuitry for controlling a current through a load. The current control circuitry comprises amplifier circuitry, reference voltage generator circuitry configured to supply a fixed reference voltage to a first input of the amplifier circuitry and an output stage comprising: a control terminal coupled to an output of the amplifier circuitry; a current input terminal configured to be coupled to the load; and a current output terminal. The current control circuitry further comprises a variable resistance coupled to the current output terminal of the output stage, and a feedback path between the current output terminal of the output stage and a second terminal of the amplifier circuitry for providing a feedback voltage to a second input of the amplifier circuitry.

FIELD OF THE INVENTION

The present disclosure relates to the field of current controlcircuitry.

BACKGROUND

A current sink is a port or circuit that accepts a negative current,e.g. a current that flows into the current sink is drained to ground.Current sink driver circuitry can be used to drive a load such as, forexample, an LED (light emitting diode). Current sink driver circuitrymay be configured to control an amount of current that flows through theload. For example, where current sink driver circuitry is used to drivean LED, the current through the LED may be controllable so as to controlthe brightness of the LED.

SUMMARY

According to a first aspect, the invention provides current controlcircuitry for controlling a current through a load, the current controlcircuitry comprising:

-   -   amplifier circuitry;    -   reference voltage generator circuitry configured to supply a        fixed reference voltage to a first input of the amplifier        circuitry;    -   an output stage comprising:        -   a control terminal coupled to an output of the amplifier            circuitry;        -   a current input terminal configured to be coupled to the            load; and        -   a current output terminal;    -   a variable resistance coupled to the current output terminal of        the output stage; and    -   a feedback path between the current output terminal of the        output stage and a second terminal of the amplifier circuitry        for providing a feedback voltage to a second input of the        amplifier circuitry.

The reference voltage generator may circuitry comprise:

-   -   a current source configured to generate a fixed current; and    -   a resistance.

The output stage may comprise a MOSFET device.

A resistance value of the variable resistance may be digitallycontrollable.

The variable resistance may comprise a resistive digital to analogueconverter (DAC).

A resistance of the resistive DAC may be based on a digital code inputto the resistive DAC.

The resistive DAC may be configured to receive a clock signal.

A change in a resistance of the resistive DAC in response to a change inthe digital code input to the resistive DAC may be synchronised to theclock signal.

The current control circuitry may further comprise a switch, operable tocouple the first and second inputs of the amplifier circuitry during achange from one digital code input to the resistive DAC to anotherdigital code input to the resistive DAC.

The current control circuitry may further comprise a current senseresistor coupled in series between the variable resistance and ground.The current sense signal is configured to generate a signal indicativeof the current through the load.

The current control circuitry may further comprise processing circuitryconfigured to adjust the operation of the current control circuitrybased on the signal indicative of the current through the load.

The processing circuitry may be configured to reduce a supply voltage orpower down the current control circuitry in response to determining thatthe current though the load exceeds a predetermined threshold.

The processing circuitry may be configured to adjust an operatingparameter of the current control circuitry to reduce an error between ameasured current through the load and a predefined current value.

For example, the processing circuitry may be operative to adjust one ormore of:

-   -   a gain of the amplifier circuitry;    -   a reference current that is used to generate the reference        voltage;    -   a resistance value of a resistance that is used to generate the        reference voltage    -   a supply voltage to a portion of the current control circuitry;        and    -   a resistance value of the variable resistance.

The current sense resistor may be of a different type than the variableresistance.

For example, the current sense resistor may be a TaN (tantalum nitride)resistor.

The current control circuitry may be configured to receive a first powersupply voltage for powering the reference voltage generator circuitryand a second power supply voltage for supplying current to the load.

The load may be a light emitting diode (LED).

According to a second aspect, the invention provides current controlcircuitry for controlling a current through a load, the current controlcircuitry comprising:

-   -   amplifier circuitry;    -   reference voltage generator circuitry configured to supply a        reference voltage to a first input of the amplifier circuitry;    -   an output stage comprising:        -   a control terminal coupled to an output of the amplifier            circuitry;        -   a current input terminal configured to be coupled to the            load; and        -   a current output terminal;    -   a feedback path between the current output terminal of the        output stage and a second terminal of the amplifier circuitry        for providing a feedback voltage to a second input of the        amplifier circuitry; and    -   a current sense resistor coupled to ground in a current path for        the load current.

According to a second aspect, the invention provides an integratedcircuit comprising current control circuitry according to the first orsecond aspect.

According to a fourth aspect, the invention provides an electronicdevice comprising an integrated circuit according to the third aspect.

The device may comprise a mobile telephone, a tablet or laptop computer,a wearable device, a gaming device, a virtual reality or augmentedreality device, for example.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the invention will now be described, strictly by way ofexample only, with reference to the accompanying drawings, of which:

FIG. 1 is a schematic diagram illustrating example current controlcircuitry;

FIG. 2 is a schematic diagram illustrating alternative example currentcontrol circuitry;

FIG. 3 is a schematic diagram illustrating example current controlcircuitry according to the present disclosure;

FIG. 4 is a schematic diagram illustrating a model of amplifiercircuitry and an output stage of the current control circuitry of FIG.3;

FIG. 5a is a schematic diagram illustrating the output stage, load andvariable resistance of the current control circuitry of FIG. 3;

FIG. 5b is a schematic diagram illustrating the effect of atransconductance of the current control circuitry of FIG. 3 with amodel;

FIG. 6 is a schematic diagram illustrating a resistive digital toanalogue converter suitable for use as a variable resistance in thecurrent control circuitry of FIG. 3;

FIG. 7 is a schematic diagram illustrating alternative example currentcontrol circuitry according to the present disclosure;

FIG. 8 is a schematic diagram illustrating further alternative examplecurrent control circuitry according to the present disclosure;

FIG. 9 is a schematic diagram illustrating further alternative examplecurrent control circuitry according to the present disclosure;

FIG. 10 is a schematic diagram illustrating further alternative examplecurrent control circuitry according to the present disclosure;

FIG. 11 is a schematic diagram illustrating an implementation of acurrent sink system;

FIG. 12 is a schematic diagram illustrating an alternativeimplementation of a current sink system;

FIG. 13 is a schematic diagram illustrating a further alternativeimplementation of a current sink system;

FIG. 14 is a schematic diagram illustrating a further implementation ofa current sink system;

FIG. 15 is a schematic diagram illustrating a further implementation ofa current sink system;

FIG. 16 is a schematic diagram illustrating an alternative to the systemof FIG. 15;

FIG. 17 is a schematic diagram illustrating a further implementation ofa current sink system; and

FIG. 18 is a schematic diagram illustrating a further implementation ofa current sink system.

DETAILED DESCRIPTION

Referring first to FIG. 1, an example of current control circuitry isshown generally at 100. The current control circuitry 100 comprisesamplifier circuitry 110 having an output that is coupled to a control(e.g. gate) terminal of an output stage 120, which in the illustratedexample is a MOSFET device.

The current control circuitry 100 further includes voltage generatorcircuitry, which in the illustrated example comprises a controllablecurrent source 130 coupled in series with a first resistance 140 betweena first supply voltage rail 140, which provides a first supply voltageVDDa and a ground (or other reference) supply rail 150. A first node 132intermediate the controllable current source 120 and the firstresistance 130 is coupled to a first (non-inverting) input terminal ofthe amplifier circuitry 110.

A load 170, which in the illustrated example is an LED, is coupledbetween a second supply voltage rail 180 and a drain terminal of theoutput stage 120. The second supply voltage rail 180 in this exampleprovides as second supply voltage VDDp.

A second resistance 190 is coupled between a source terminal of theoutput stage 120 and the ground (or other reference) supply rail 150. Asecond node 192 intermediate the source terminal of the output stage andthe second resistance 190 is coupled to a second (inverting) input ofthe amplifier circuitry 110. Thus a feedback path is provided betweenthe second node 192 and the second input of the amplifier circuitry 110,by means of which a voltage Vfbck that develops at the second node 192can be fed back to the amplifier circuitry 110.

On startup of the current control circuitry 100, the controllablecurrent source generates a current iref, which causes a voltage Vref todevelop at the first node 132. The voltage Vref is thus supplied to thenon-inverting input of the amplifier circuitry 110 as a referencevoltage. The output stage 120 is initially switched off, so no currentflows through the load 170, and the voltage Vfbck is lower than thereference voltage Vref. The amplifier circuitry 110 thus outputs apositive voltage to the control (gate) terminal of the output stage as abias voltage, to drive the output stage. Consequently a current iloadflows through the load and the second resistance 190, and the voltageVfback develops at the second node 192.

The feedback arrangement of the amplifier circuitry 110 acts to minimiseany difference between the reference voltage Vref and the voltage Vfbck,by causing the amplifier circuitry 110 to adjust its output voltagewhich, as discussed above, is received by the control terminal of theoutput stage 120 as its bias voltage. Adjusting the bias voltage to thecontrol terminal of the output device 120 changes the load currentiload. Thus the amplifier circuitry 110 is operative to maintain aconstant load current iload for a given reference current iref.

In order to change the load current iload, the reference current iref isadjusted, by providing an appropriate control input (e.g. a controlsignal) to the controllable current source 130.

Increasing the reference current iref causes the reference voltage Vrefat the first input of the amplifier circuitry 110 to increase, which inturn causes the voltage at the output of the amplifier circuitry 110 toincrease (as the reference voltage Vref at the non-inverting input ofthe amplifier circuitry is now greater than the voltage Vfbck at theinverting input of the amplifier circuitry 110). Thus the bias voltagesupplied to the control terminal of the output stage 120 increases,thereby increasing the load current iload.

Conversely, decreasing the reference current iref causes the referencevoltage Vref at the first input of the amplifier circuitry 110 todecrease, which in turn causes the voltage at the output of theamplifier circuitry 110 to decrease (as the reference voltage Vref atthe non-inverting input of the amplifier circuitry is now smaller thanthe voltage Vfbck at the inverting input of the amplifier circuitry110). Thus the bias voltage supplied to the control terminal of theoutput stage 120 decreases, thereby decreasing the load current iload.

As will be appreciated, the rate at which the load current iload can bechanged in the current control circuitry 100 (i.e. diload/dt) is limitedby the bandwidth of the feedback loop of the amplifier circuitry 110,and thus the current control circuitry 100 may not be suited toapplications in which a high rate of change of the load current iload isrequired.

FIG. 2 shows an alternative example of current control circuitry. Thecurrent control circuitry, shown generally at 200 in FIG. 2, comprisesamplifier circuitry 210 having an output that is coupled to control(e.g. gate) terminals of matched (in terms of device properties such aschannel width and length) first and second MOSFET devices 220, 230.Source terminals of the first and second MOSFET devices are coupled to aground (or other reference) rail 240. Thus the first and second MOSFETdevices 220, 230 are arranged as a current mirror.

A controllable current source 250 is coupled between a first supplyvoltage rail 260 and a drain terminal of the first MOSFET device 220.The first supply voltage rail 260 in this example provides a firstsupply voltage VDDa. A node 252 intermediate the controllable currentsource 250 and the drain terminal of the first MOSFET device 220 iscoupled to a first (non-inverting) input of the amplifier circuitry 210.

A load 270, which in the illustrated example is an LED, is coupledbetween a second supply voltage rail 280 and a drain terminal of thesecond MOSFET device 230. The second supply voltage rail 280 in thisexample provides a second supply voltage VDDp. A node 272 intermediatethe load 270 and the drain terminal of the second MOSFET device 230 iscoupled to a second (inverting) input of the amplifier circuitry 210.Thus a feedback path is provided between the second node 272 and thesecond input of the amplifier circuitry 210, by means of which a voltageVfbck that develops at the second node 272 can be fed back to theamplifier circuitry 210.

On startup of the current control circuitry 200, the controllablecurrent source 250 generates a current iref, which causes a voltage Vrefto develop at the first node 252. The voltage Vref is thus supplied tothe non-inverting input of the amplifier circuitry 210 as a referencevoltage. Assuming that the voltage Vfbck is lower than the referencevoltage Vref, the amplifier circuitry 210 thus outputs a positive biasvoltage to the control (gate) terminal of the first and second MOSFETdevices 220, 230, to drive the first and second MOSFET devices 220, 230.Consequently a current iload, which is a copy of the current iref (dueto the current mirror arrangement of the first and second MOSFETs 220,230) flows through the load 270, and the voltage Vfback develops at thesecond node 272.

The feedback arrangement of the amplifier circuitry 210 acts to minimiseany difference between the reference voltage Vref and the voltage Vfbck,by causing the amplifier circuitry 210 to adjust its output voltagewhich, as discussed above, is received as a bias voltage by the controlterminals of the first and second MOSFET devices 220, 230.

In order to change the load current iload, the reference current iref isadjusted, by providing an appropriate control input (e.g. a controlsignal) to the controllable current source 250.

Increasing the reference current iref causes the reference voltage Vrefat the first input of the amplifier circuitry 210 to increase, which inturn causes the voltage at the output of the amplifier circuitry 210 toincrease (as the reference voltage Vref at the non-inverting input ofthe amplifier circuitry is now greater than the voltage Vfbck at theinverting input of the amplifier circuitry 110). Thus the bias voltagesupplied to the control terminals of the first and second MOSFET devices220, 230 increases, thereby increasing the load current iload.

Conversely, decreasing the reference current iref causes the referencevoltage Vref at the first input of the amplifier circuitry 210 todecrease, which in turn causes the voltage at the output of theamplifier circuitry 210 to decrease (as the reference voltage Vref atthe non-inverting input of the amplifier circuitry is now smaller thanthe voltage Vfbck at the inverting input of the amplifier circuitry210). Thus the bias voltage supplied to the control terminals first andsecond MOSFET devices 220, 230 decreases, thereby decreasing the loadcurrent iload.

In a practical implementation of the circuitry 200 the matching betweenthe first and second MOSFET devices 220, 230 is typically not perfect.There may be a difference of as much as 10% between the relevantcharacteristics of the first and second MOSFET devices 220, 230. As willbe appreciated, such imperfect matching will adversely affect theaccuracy with which the load current iload can be controlled.Additionally, the rate at which the load current iload can be changed inthe current control circuitry 100 (i.e. diload/dt) is limited by thebandwidth of the feedback loop of the amplifier circuitry 210. Thus thecurrent control circuitry 200 may not be suited to applications in whicha high degree of accuracy of the load current iload, or a high rate ofchange of the load current iload, is required.

FIG. 3 is a schematic representation of an example of current controlcircuitry according to the present disclosure.

The circuitry, shown generally at 300 in FIG. 3, comprises amplifiercircuitry 310 having an output that is coupled to a control (e.g. gate)terminal of an output stage 320, which in the illustrated example is aMOSFET device.

The current control circuitry 300 further includes voltage generatorcircuitry, which in the illustrated example comprises a current source330 coupled in series with a first resistance 340 between a first supplyvoltage rail 360, which provides a first supply voltage VDDa and areference supply rail 350, which in this example is coupled to ground.In contrast to the circuitry 100 of FIG. 1, the current source 330 isconfigured to output generate a fixed current.

A first node 332 intermediate the controllable current source 320 andthe first resistance 330 is coupled to a first (non-inverting) inputterminal of the amplifier circuitry 310.

A load 370, which in the illustrated example is an LED, is coupledbetween a second supply voltage rail 380 and a drain terminal of theoutput stage 320. The second supply voltage rail 380 in this exampleprovides as second supply voltage VDDp.

A variable resistance 390 is coupled between a source terminal of theoutput stage 320 and the reference supply rail 350. A second node 392intermediate the source terminal of the output stage and the variableresistance 390 is coupled to a second (inverting) input of the amplifiercircuitry 310. Thus a feedback path is provided between the second node392 and the second input of the amplifier circuitry 310, by means ofwhich a voltage Vfbck that develops at the second node 392 can be fedback to the amplifier circuitry 310.

On startup of the current control circuitry 300, the current sourcegenerates a current iref, which causes a voltage Vref to develop at thefirst node 332. The voltage Vref is thus supplied to the non-invertinginput of the amplifier circuitry 310 as a reference voltage. The outputstage 320 is initially switched off, so no current flows through theload 370 and the voltage Vfbck is lower than the reference voltage Vref.The amplifier circuitry 310 thus outputs a positive voltage to thecontrol (gate) terminal of the output stage 320 as a bias voltage, todrive the output stage 320. Consequently a current iload flows throughthe load 370 and the variable resistance 390, and the voltage Vfbackdevelops at the second node 392.

The feedback arrangement of the amplifier circuitry 310 acts to minimiseany difference between the reference voltage Vref and the voltage Vfbck,by causing the amplifier circuitry 310 to adjust its output voltagewhich, as discussed above, is received by the control terminal of theoutput stage 320 as its bias voltage.

In order to change the load current iload, the value of the variableresistance 390 is adjusted, by providing an appropriate control input(e.g. a control signal) to the variable resistance 390.

Reducing the value of the variable resistance 390 causes aninstantaneous increase in the load current iload. Because the biasvoltage to the output stage 320 cannot immediately change in response tothe increase in the load current iload, the reduction in the loadcurrent iload gives rise to a relatively large voltage decrease in thevoltage Vfbck at the second node 392.

As the reference voltage Vref at the non-inverting input of theamplifier circuitry 310 is constant, a large difference now existsbetween the voltage Vref at the inverting input of the amplifiercircuitry and the voltage Vfbck at the non-inverting input of theamplifier circuitry 310. The amplifier circuitry 310 responds quickly tothis large difference by increasing the voltage at its output, thusincreasing the bias voltage to the output stage 320. The voltage Vfbckat the second node 392 (and thus at the inverting input of the amplifiercircuitry 310) thus increases to compensate for the initial large dropin voltage at the second node 392 and therefore equalise the voltages atthe inputs to the amplifier circuitry 310.

Increasing the value of the variable resistance 390 causes aninstantaneous decrease in the load current iload. Again, because thebias voltage to the output stage 320 cannot immediately change inresponse to the increase in the load current iload, the decrease in theload current iload gives rise to a relatively large increase in thevoltage Vfbck at the second node 392. The amplifier circuitry 310responds quickly to this large difference by reducing the voltage at itsoutput, thus decreasing the bias voltage to the output stage 320. Thevoltage Vfbck at the second node 392 (and thus at the inverting input ofthe amplifier circuitry 310) is thus reduced to compensate for theinitial increase in voltage at the second node 392 and thereforeequalise the voltages at the inputs to the amplifier circuitry 310.

Thus, in the circuitry 300 of FIG. 3 the rate of change diload/dt of theload current iload is dependent upon the rate at which the value of thevariable resistance 390 can be changed. If a suitable variableresistance 390 is employed a much higher rate of change diload/dt of theload current iload than in the circuitry 100 of FIG. 1.

Further, the accuracy of the change in the load current iload isdependent upon the accuracy of the variable resistance 390, rather thanbeing dependent on a matching accuracy between two MOSFET devices, as inthe circuitry 200 of FIG. 2. As the accuracy of a resistance isgenerally much greater than the matching accuracy of two differentMOSFET devices, the circuitry 300 can provide greater accuracy in theload current iload than the circuitry 200 of FIG. 2. For example, thematching accuracy between two MOSFET devices may be around 10%, whereasthe value of an individual resistor may be within 1% of its nominal orrated value.

Thus the circuitry 300 is able to control the load current iload withgreater speed and greater accuracy than the circuitry 100, 200 of FIGS.1 and 2.

Another factor that contributes to the improved rate of change of theload current iload in the circuitry 300 will now be described withreference to FIG. 4.

The amplifier 310 may be configured as a fixed gain amplifier whichamplifies the difference between the signals at its inputs. The outputstage 320 acts as an attenuator. Thus the amplifier 310 and output stage320 can be modelled as a fixed gain amplifier 410 having an output thatis coupled to an input of an attenuator 420, as shown in FIG. 4.

The attenuation in the output stage 320 arises as a result of thetransconductance gm of the output stage 320. This can be modelled as aresistance Rgm 510 in series with the variable resistance 370, as shownin FIG. 5b (FIG. 5a shows the output stage 320, load 370 and variableresistance 390 for comparison). The combination of the resistance Rgm510 and the variable resistance 390 effectively forms a voltage divider,and thus the attenuation that arises due to the resistance Rgm 510 canbe calculated as:

${{Attenuation} = \frac{Rv}{{Rv} + {Rgm}}},$

Where Rv is the resistance value of the variable resistance 390.

As the resistance Rgm is the reciprocal of the transconductance gm

$\left( {{i.e.\mspace{14mu}{Rgm}} = \frac{1}{gm}} \right),$

and the transconductance gm is a function of the current flowing throughthe output stage 320

$\left( {{e.g.\mspace{14mu}{gm}} = \sqrt{k \cdot {iload}}} \right),$

then it follows that

${Attenuation} = {\frac{Rv}{{Rv} + \frac{1}{\sqrt{k \cdot {iload}}}}.}$

Thus, as the load current iload decreases as a result an increase in theresistance value Rv of the variable resistance 390, the attenuationtends towards 1.

This is because for a given change in Rv, the resulting change in Rgmwill be much smaller, since

${Rgm} = {\frac{1}{\sqrt{k \cdot \frac{Vfbck}{Rv}}}.}$

Thus, for a small change in the load current iload (resulting from asmall change in the value of the variable resistance 390), theattenuation by the output stage 320 has only a very small effect on thevoltage Vfbck at the second node 392. The attenuation by the outputstage 320 thus does not significantly reduce the loop bandwidth of aloop 315 formed by the amplifier circuitry 310, the output stage 320 andthe associated feedback path between the node 392 and the invertinginput of the amplifier circuitry 310, and therefore does notsignificantly affect the speed with which the loop 315 compensates forthe change in load current iload.

The variable resistance 390 may be implemented in a number of ways. Forexample, the variable resistance 390 may be implemented as a resistivedigital to analogue converter (DAC), using switched resistances, asillustrated in FIG. 6.

FIG. 6 shows an example of a four-bit resistive DAC 600, which comprisesfirst to fourth banks 610-640 of switched resistances.

The first bank 610 comprises a first resistance 612 of value R coupledin series with a first switch 614 between a first rail 650 that iscoupled to the second node 392 of the circuitry 300 and a second rail660 that is coupled to the reference supply rail 350 of the circuitry300. Although for clarity the first resistance 612 is shown in FIG. 6 asa single resistor, it will be appreciated by those skilled in the artthat the first resistance 612 could be made up of a number of separateresistances coupled in series or parallel in order to achieve theresistance value R.

The second bank 620 comprises a second resistance 622 of value R/2coupled in series with a second switch 624 between the first rail 650and the second rail 660. Again, for clarity the second resistance 622 isshown in FIG. 6 as a single resistor, but it will be appreciated bythose skilled in the art that the second resistance 622 could be made upof a number of separate resistances coupled in series or parallel inorder to achieve the resistance value R/2.

The third bank 630 comprises a third resistance 632 of value R/4 coupledin series with a third switch 634 between the first rail 650 and thesecond rail 660. As before, for clarity the third resistance 632 isshown in FIG. 6 as a single resistor, but it will be appreciated bythose skilled in the art that the third resistance 632 could be made upof a number of separate resistances coupled in series or parallel inorder to achieve the resistance value R/4.

The fourth bank 640 comprises a fourth resistance 642 of value R/8coupled in series with a fourth switch 644 between the first rail 650and the second rail 660. Again, for clarity the fourth resistance 642 isshown in FIG. 6 as a single resistor, but it will be appreciated bythose skilled in the art that the fourth resistance 642 could be made upof a number of separate resistances coupled in series or parallel inorder to achieve the resistance value R/8.

The resistance value of the variable resistance 390 can be adjusted byselectively opening and closing the switches 614-644 in accordance with,in this example, a four-bit input digital word or code, and this allowsthe load current iload to be adjusted in increments equal to 1 LSB(least significant bit) of the resistive DAC. The resistance value ofthe variable resistance is therefore based on the input digital word orcode.

Thus for an input digital word of value 0001, the first switch 614 wouldbe closed and the second, third and fourth switches 624-644 would beopen. The resistance value of the variable resistor 390 would thus beequal to R, and the load current iload would be equal to Vref/R (becausein steady state operation of the loop 315 Vref=Vfbck).

For an input digital word of value 0010, the second switch 624 would beclosed and the first, third and fourth switches 614, 634, 644 would beopen. The resistance value of the variable resistor 390 would thus beequal to R/2, and the load current iload would be equal to 2Vref/R.

For an input digital word of value 0011, the first and second switches614, 624 would be closed and the third and fourth switches 634, 644would be open. The resistance value of the variable resistor 390 wouldthus be equal to the parallel combination of R and R/2, i.e. R/3, andthe load current iload would be equal to 3Vref/R.

Thus the load current iload can be adjusted in increments of Vref/R.Because the resistances 612-642 can be manufactured with very precisetolerances (whether they are provided as single resistors or serialand/or parallel combinations of multiple resistors), the load currentincrements can be very accurate.

The use of a resistive DAC 600 as the variable resistance 390 permitsdigital control of the load current iload. Typically the resistive DAC600 is configured to receive a clock signal, and so any change in theresistance of the resistive DAC in response to a change in an inputdigital word or code value is synchronised to the clock signal. Thus therate of change of the load current iload is dependent upon the clockfrequency, in the sense that only one load current increment can beperformed per clock cycle. Accordingly, a high rate of change of theload current (i.e. a high diload/dt) can be achieved by using ahigh-frequency clock.

One issue that can arise when using a resistive DAC to provide thevariable resistance 390 is that the resistive DAC can temporarily adopta high impedance state during transitions between input codes (i.e.during a change from one input digital code value to a different inputdigital code value), as all of the switches 614-644 may be opensimultaneously during the transition, before the appropriate ones of theswitches 614-644 are closed to achieve the desired resistance value,thus effectively making the variable resistance 390 open-circuittemporarily.

As a result of this temporary high impedance state, a voltage spike mayappear at the second node 392. This causes the voltage Vfbck at theinverting input of the amplifier circuitry 310 to be higher than thereference voltage Vreg at the non-inverting input. The voltage at theoutput of the amplifier circuitry 310 is thus reduced, reducing the biasvoltage to the output stage 320 and thus reducing the load currentiload.

Where the change of input code is intended to reduce the value of thevariable resistance (i.e. the value of the input code increases) andthus increase the load current iload, this momentary reduction in theload current iload can increase the time required to reach the desiredload current iload, because the load current iload has to increase froma lower starting value to the desired value.

FIG. 7 is a schematic representation of an example of current controlcircuitry according to the present disclosure that can mitigate thisproblem. The current control circuitry, shown generally at 700 in FIG.7, includes a number of features in common with the circuitry 300 ofFIG. 3. Such common features are denoted by common reference numerals inFIGS. 3 and 7 and will not be described again here, for the sake ofbrevity.

The current control circuitry 700 differs from the circuitry 300 of FIG.3 in that it includes a switch 710 (e.g. a transistor) that is operableto couple (i.e. short-circuit) the first and second inputs of theamplifier circuitry 310. The switch 710 can be actuated during an inputcode transition (i.e. a change from one input digital code to aresistive DAC implementation of the variable resistance 390) to ensurethat the same voltage is received at both of the inputs of the amplifiercircuitry 310 during the input code transition, such that the voltageoutput by the amplifier circuitry 310 does not change during the inputcode transition, thereby avoiding the momentary reduction in the loadcurrent iload described above. Once the input code has settled to thedesired value, the switch 710 can be opened, allowing the amplifiercircuitry 310 to respond to any difference between the voltages at itsinputs that may arise as a result of the change in the input code asdescribed above with reference to FIG. 3.

FIG. 8 is a schematic representation of a further example of currentcontrol circuitry according to the present disclosure. The currentcontrol circuitry, shown generally at 800 in FIG. 8, includes a numberof features in common with the circuitry 300 of FIG. 3. Such commonfeatures are denoted by common reference numerals in FIGS. 3 and 8 andwill not be described again here, for the sake of brevity.

The current control circuitry 800 differs from the circuitry 300 of FIG.3 in that the reference supply rail 350 is not coupled directly toground, but is instead coupled to a first terminal of a sense resistor810. A second terminal of the sense resistor 810 is coupled to ground.Thus the current sense resistor 810 is coupled between the variableresistance 390 and ground, and is thus coupled to ground in a currentpath for the load current iload.

In some examples the current sense resistor 810 may be of a differenttype than the variable resistance 390. For example, the variableresistance may be made up of a plurality of polysilicon (sometime alsoreferred to simply as poly) resistors, whereas the current senseresistor 810 may be a TaN (tantalum nitride) resistor. As TaN resistorsgenerally have low temperature coefficients, the use of a TaN resistoras the current sense resistor 810 minimises or at least reduces theeffect (in comparison to polysilicon resistors, for example) oftemperature changes on the accuracy of load current measurement.

The circuitry 800 operates in the same way as the circuitry 300described above. The presence of the sense resistor 810 does not changethe operation of the circuitry, as the amplifier circuitry 310 stillreceives a constant reference voltage Vref at its non-inverting input,which is compared to a voltage Vfbck, received at the inverting input ofthe amplifier circuitry 310, that is dependent upon the resistance valueof the variable resistance 390.

However, the sense resistor 810 permits the current iload to bemeasured. Thus a voltage Vsense across the sense resistor 810 can bemeasured, and an indication of the load current iload can be determinedbased on the measured voltage Vsense and the resistance value of thesense resistor 810.

Alternatively, an analogue to digital converter (ADC) 820 may be coupledto the first terminal of the sense resistor 810 and configured toconvert the analogue voltage Vsense across the sense resistor 810 to adigital output signal that provides an indication of the load currentiload.

The indication of the load current iload (whether determined based onthe measured voltage Vsense or provided as a digital output of the ADC820) may be used by processing circuitry 830 to ensure that the loadcurrent iload is within specified limits and thus that the load 370 isnot drawing excessive current.

The processing circuitry 830 may be, for example, a dedicatedmicroprocessor, microcontroller, state machine or the like, or mayalliteratively be a processing resource such as a main processor or anapplications processor of a host device that incorporates the circuitry800.

If the processing circuitry 830 determines that the load current iloadis outside of the specified limits (e.g. if the processing circuitry 830determines that the load current iload exceeds a predeterminedthreshold, and thus that the load 370 may be drawing excessive current)the processing circuitry 830 may adjust the operation of the circuitry800, e.g. by powering down the circuitry 800 or reducing the secondsupply voltage VDDp.

Additionally or alternatively, the indication of the load current iload(whether determined based on the measured voltage Vsense or provided asa digital output of the ADC 820) may be used as a feedback signal tomodify or adjust one or more operating parameters of the circuitry 800so as to improve the accuracy of the load current iload. For example,the processing circuitry 830 may compare the indication of the loadcurrent iload to a predefined current value representing a desired orideal value of the load current iload, and may adjust one or moreparameters of the circuitry 800, e.g. the gain of the amplifiercircuitry 310, the second supply voltage VDDp, the reference currentiref, the resistance value of the resistance 340, or the resistancevalue of the variable resistance 390, in order to reduce or minimise anyerror between the measured iload current (as sensed by the senseresistor 810) and the predefined current value representing the desiredor ideal value of the load current iload.

A similar arrangement of a sense resistor 810, processing circuitry 830and (optionally) an ADC 820 may be added to the circuitry 100 of FIG. 1(as illustrated generally at 900 in FIG. 9) or to the circuitry 200 ofFIG. 2 (as illustrated generally at 1000 in FIG. 10).

The arrangements 900, 1000 illustrated in FIGS. 9 and 10, in which acurrent sense resistor 810 is coupled to ground in a current path forthe load current iload, permit the load current iload to be measured andthe operation of the circuitry 900, 1000 to be adjusted as describedabove in the event that the processing circuitry 830 determines that theload current iload is outside of the specified limits.

The arrangements 900, 1000 illustrated in FIGS. 9 and 10 also permit orone or more operating parameters of the circuitry 100, 200 (e.g. thecurrent iref, a gain of the amplifier circuitry 110, 210, a resistancevalue of the resistance 140, or a resistance value of the resistance190) to be adjusted or modified in response to an indication of the loadcurrent in order to improve the accuracy of the load current iload, asdescribed above.

In the foregoing description and the accompanying drawings, an LED isused as an example of the load 370. However, it will be appreciated bythose of ordinary skill in the art that the circuitry 300, 700, 800 ofthe present disclosure is equally suitable for driving other loads.

Further, although the example circuitry 300, 700, 800 described aboveincludes first and second supply voltages VDDa, VDDp, it will beappreciated by those of ordinary skill in the art that the circuitry300, 700, 800 can also operate from a single supply voltage.

As will be apparent from the foregoing description, the current controlcircuitry of the present disclosure permits fast and highly accuratecontrol of a current though a load.

The use of the variable resistance 390 permits the load current iload tobe controlled directly, and thus the speed with which the load currentcan change is not dependent upon the loop bandwidth of a loop 115, 215made up of an amplifier, an output stage and an associated feedbackpath, as in the circuitry 100, 200 shown in FIGS. 1 and 2. Inembodiments in which the variable resistance is implemented using aresistive DAC, the rate of change of the load current is dependent uponthe frequency of a clock to which the resistive DAC is synchronised.Thus a high rate of change of load current can be achieved if a highfrequency clock is employed.

Further, as the current control circuitry of the present disclosure doesnot use matched MOSFET devices, but instead uses only a variableresistance to control the load current, the accuracy of the load currentcan be increased, in comparison to the accuracy that is achievable usingthe circuitry 200 of FIG. 2.

Embodiments may be implemented as an integrated circuit which in someexamples could be a codec or audio DSP or similar. Embodiments may beincorporated in an electronic device, which may for example be aportable device and/or a device operable with battery power. The devicecould be a communication device such as a mobile telephone or smartphoneor similar. The device could be a computing device such as a notebook,laptop or tablet computing device. The device could be a wearable devicesuch as a smartwatch. The device could be a device with voice control oractivation functionality such as a smart speaker. In some instances thedevice could be an accessory device such as a headset, headphones,earphones, earbuds or the like to be used with some other product. Insome instances the device could be a gaming device such as a gamesconsole, or a virtual reality (VR) or augmented reality (AR) device suchas a VR or AR headset, spectacles or the like.

The skilled person will recognise that some aspects of theabove-described apparatus and methods, for example the discovery andconfiguration methods may be embodied as processor control code, forexample on a non-volatile carrier medium such as a disk, CD- or DVD-ROM,programmed memory such as read only memory (Firmware), or on a datacarrier such as an optical or electrical signal carrier. For manyapplications, embodiments will be implemented on a DSP (Digital SignalProcessor), ASIC (Application Specific Integrated Circuit) or FPGA(Field Programmable Gate Array). Thus the code may comprise conventionalprogram code or microcode or, for example code for setting up orcontrolling an ASIC or FPGA. The code may also comprise code fordynamically configuring re-configurable apparatus such asre-programmable logic gate arrays. Similarly the code may comprise codefor a hardware description language such as Verilog™ or VHDL (Very highspeed integrated circuit Hardware Description Language). As the skilledperson will appreciate, the code may be distributed between a pluralityof coupled components in communication with one another. Whereappropriate, the embodiments may also be implemented using code runningon a field-(re)programmable analogue array or similar device in order toconfigure analogue hardware.

Further Example Embodiments of the Present Disclosure

The following discussion provides further disclosure relating to currentsinks and current sink drivers.

The description below sets forth example embodiments according to thisdisclosure. Further example embodiments and implementations will beapparent to those having ordinary skill in the art. Further, thosehaving ordinary skill in the art will recognize that various equivalenttechniques may be applied in lieu of, or in conjunction with, theembodiments discussed below, and all such equivalents should be deemedas being encompassed by the present disclosure.

Description

A current sink is a port or circuit that accepts a negative current,e.g. current into the circuit which is drained to ground. Current sinkdrivers can be used to drive circuit elements, e.g. LEDs. The followingdescribes examples of current sink designs providing high accuracycombined with low voltage headroom requirements.

With reference to FIG. 11, the current sink A receives a referencecurrent Idac, e.g. from a current DAC, and provides a sink for theoutput current Isnk, where Isnk=A·Idac.

With reference to FIG. 12, a first implementation of a current sourcesystem is shown. A voltage to current (V2I) converter is used togenerate a reference current for a current DAC from a bandgap voltage.The V2I converter may be provided with a resistor, such as a TaNresistor, which can be trimmed for accurate current output, e.g. byremoving the TaN process variation, V2I converter amplifier input offsetand/or initial bandgap voltage error.

The reference current is provided to a current DAC (or IDAC) to generatethe reference current Idac for the current sink. It will be understoodthat multiple current sinks may be provided within a single device, toprovide multiple output sink currents, e.g. in the above figure tenseparate output currents are provided as ISNK<9:0>. It will beunderstood that the V2I converter and IDAC may be used to generateseparate Idac reference currents for each current sink

The current sink comprises an amplifier and multiple switch devices,e.g. MOSFETS as illustrated above.

The Idac reference current drives the reference device (1) of thecurrent sink. The Idac reference device current is mirrored and scaledusing an output device (100) to provide the output current of thecurrent sink, e.g. the above-illustrated implementation provides anoutput current of 100 times the Idac reference current. An amplifier isused to force the reference device and output device to have the sameVds voltage.

The sink is further provided with a device to provide overvoltageprotection, with an appropriately rated (e.g. 5 Volt) DMOS (diffusedmetal-oxide semiconductor) such as an EDMOS or an LDMOS used to protectthe amplifier and the connected devices. The protection device is usedto protect the amplifier inverting input and the output device. Theprotection device is normally switched on, and has a low voltage DC biasat its gate. The protection device acts as a voltage follower, itssource being at a lower voltage than its gate. As EDMOS and LDMOSdevices can withstand high drain to gate voltages, potentially damagingvoltages at the current sink path can be conducted away from theamplifier inverting input and the output device by the protectiondevice, thus protecting the amplifier and the output device.

The current sink is further provided with a 1-ohm current sense resistoron the Isnk path, to allow for detection of the current through thesink.

A second implementation of current sink is illustrated in FIG. 13, basedon the implementation of FIG. 12, with the current sense resistorsground-referenced to ease current sense path design. In the above and insubsequent figures the V2I converter and IDAC are not shown, and arereplaced by the device above named IDAC, which will be understood asproviding a reference current for the current sink, for example usingthe approach as shown in FIG. 12.

In situations where the use of a protection device is not possible,adding an additional switch in series with the resistors is notdesirable as it will further degrade the available headroom. In such ascenario, two output devices are used as shown above, a large device forhigh current range (e.g. approximately 20-50 mA) and a second deviceapprox. 60% smaller for low current range (e.g. 0-20 mA). Switches areused to select which output device the amplifier drives.

As two different output devices are being used, the design must alsoswitch between two different reference devices. To match the referencesto the output device a TaN resistor of ×100 the sense resistor is addedin the source of the references.

FIG. 14 illustrates a simplification of the system of FIG. 13, whereinthe range selection is provided using a MOSFET switch arranged to bypassa resistance on the output path. This approach reduces the total numberof devices and circuit area required.

FIG. 15 illustrates a further implementation of a current sink.

In the FIG. 15 implementation, the current sense resistor isground-referenced, to ease the design of the current sense path. AnEDMOS device is used as the sink, thereby removing the need for aseparate protection device. One amplifier input is taken from the IDACinput, with the other input to the amplifier provided via the sink path,after the EDMOS device, so that the requirement for current mirrormatching is removed, and the amplifier doesn't connect to the outputpin.

As a result, the implementation of FIG. 15 reduces the headroomrequirements relative to the implementation of FIG. 12.

A further implementation of current sink is illustrated in FIG. 16.

In this implementation, the IDAC current defines the sink current, aspreviously described. The IDAC current flows through a first resistordefining the reference voltage for the amplifier, wherein the amplifieroutputs a voltage to define the sink current through the sink path,based on feedback voltage taken from the sink path across a sink pathresistor.

The inputs to the amplifier are referenced to ground (i.e. the amplifierdoes not allow for an input swing to the level of the supply voltage)—asa result the amplifier has a relatively narrow common mode input range,thereby allowing a conventional PMOS input stage to be used. As theamplifier is ground-referenced, no relatively high voltages will beinput to the amplifier. No additional high voltage protection devicesare required, just one high voltage power device. In addition, theheadroom requirements for the IDAC current reference are relaxed.

A separate resistor to ground can be used to sense the current throughthe sink device, preferably a TaN resistor. As a result, the currentgeneration stage and the sensing stages are independent of each other,allowing for improved robustness of design.

An alternative to the system of FIG. 16 is shown in FIG. 17. In such aconfiguration, the output resistor (referenced as P+ poly resistor inthe above, but it will be understood that other resistor types may beused) is configured as the variable element in the form of an RDAC,instead of the variable current source of FIG. 16.

By moving the DAC to a resistor on the output, accordingly this mayprovide advantages in reducing settling time of the device, as thevariability introduced by the RDAC does not significantly impact theslew rate of the system. The settling may be optimized over the fullcurrent range of the device, or the gain may be kept constant over thefull current range of the device.

A higher resistance of the RDAC may be used at lower currents, whichmaintains a high gain of the loop and reduced gate movement for steps inthe current, e.g. a 50 uA step. A relatively reduced resistance of theRDAC may be used at high currents. Accordingly, the available headroomfor the device at high current levels may be relatively unchangedcompared to the FIG. 16 implementation. The RDAC may be designed for ˜1%resistor matching accuracy.

As a result of using this approach, the input voltage to the amplifieris fixed. Accordingly, the input offset is not signal dependent, and isless dominant on accuracy. The amplifier is effectively bypassed whenthe resistance of the RDAC is adjusted, and the output stage adjusts toaccommodate the change in current through the sink path, such that theamplifier can slew more slowly as the output current is already close tothe required level.

A further implementation of current sink is illustrated in FIG. 18. Inthis implementation, the current sink is provided with two outputstages, to allow for a wider range of sink current to be accommodated.An output stage will be understood as meaning a current path or a seriesof scaled devices which are provided in parallel, or configured to allowfor a number of different sink current paths to be defined, thedifferent sink current paths arranged to sink different levels ofcurrent.

A pair of parallel current sink paths are provided between ISNK andground, where the output of the amplifier as well as the feedback inputto the amplifier are multiplexed to allow the selection of differentcurrent sink paths.

The switch device and/or resistances provided in each path are scaled toallow for the sinking of different current levels in the differentpaths. In the above figure, the device and resistances in a first pathare scaled to allow a first level of sink current, e.g. between 0-20 mA,while the device and resistances in the second path are scaled to allowa relatively larger level of current flow, e.g. between approximately20-50 mA.

In addition, the current sink device is provided with two referenceinput stages, wherein the reference current received from the IDAC ismultiplexed between two different reference resistances, depending onthe output stage used. The reference amplifier input is multiplexedaccordingly.

It will be understood that the switching of the above-described currentsink paths and current reference paths may be selected based on a targetcurrent to be output, and may be controlled using a separate controlmodule (not shown), e.g. a DSP module, an applications processor, ordevice CPU.

The ranges can be selected to maintain the same reference current andtherefore preserve any error formed by the amplifier offset voltage. Inaddition, the implementation shown above avoids adding switches in thecurrent paths, thereby maintaining a straight ratio for resistors to setthe ratio of input to output current. The use of two output stages forthe current sink can allow for an improved balance between deviceheadroom and the magnitude of feedback voltage. It will be understoodthat while two output stages are illustrated in FIG. 16, a current sinkdevice may be provided having a plurality of output stages to allow fora wider range of device sink current.

The above-described current sink implementations provide high-accuracycurrent sinks having relatively low voltage headroom requirements. Thecurrent sinks may be used as current sink drivers for any suitablesystem, e.g. as a current sink LED driver.

Embodiments of the above-described systems may be implemented in a hostdevice, especially a portable and/or battery powered host device such asa mobile computing device for example a laptop or tablet computer, awearable device, a games console, a remote control device, a homeautomation controller or a domestic appliance including a domestictemperature or lighting control system, a toy, a machine such as arobot, an audio player, a video player, or a mobile telephone forexample a smartphone. There is further provided a host deviceincorporating the above-described system. There is further provided acontrol method for a system as described above.

It should be noted that the above-mentioned embodiments illustraterather than limit the invention, and that those skilled in the art willbe able to design many alternative embodiments without departing fromthe scope of the appended claims. The word “comprising” does not excludethe presence of elements or steps other than those listed in a claim,“a” or “an” does not exclude a plurality, and a single feature or otherunit may fulfil the functions of several units recited in the claims.Any reference numerals or labels in the claims shall not be construed soas to limit their scope.

It should be understood—especially by those having ordinary skill in theart with the benefit of this disclosure—that the various operationsdescribed herein, particularly in connection with the figures, may beimplemented by other circuitry or other hardware components. The orderin which each operation of a given method is performed may be changed,and various elements of the systems illustrated herein may be added,reordered, combined, omitted, modified, etc. It is intended that thisdisclosure embrace all such modifications and changes and, accordingly,the above description should be regarded in an illustrative rather thana restrictive sense.

Similarly, although this disclosure makes reference to specificembodiments, certain modifications and changes can be made to thoseembodiments without departing from the scope and coverage of thisdisclosure. Moreover, any benefits, advantages, or solutions to problemsthat are described herein with regard to specific embodiments are notintended to be construed as a critical, required, or essential featureor element.

Further embodiments likewise, with the benefit of this disclosure, willbe apparent to those having ordinary skill in the art, and suchembodiments should be deemed as being encompassed herein.

Aspects of the system may be defined by the following numberedstatements:1. There is provided a current sink driver comprising:

An input to receive a reference current via a current reference path;

An output to define a sink current to flow via a current sink path toground;

An amplifier having first and second input terminals to generate a sinkpath voltage, the sink path voltage used to define the sink current;

Wherein the first amplifier input is coupled with the current referencepath, the second amplifier input being feedback connected with thecurrent sink path, and

wherein the first and second inputs of the amplifier are connected toground via respective resistances.

Providing such an amplifier configuration having inputs referenced toground minimizes amplifier voltage swing.2. Preferably, the current sink driver comprises a reference pathresistor, the first input of the amplifier connected to ground throughthe reference path resistor.3. Preferably, the current sink driver comprises a sink path resistor,the second input of the amplifier connected to ground through the sinkpath resistor.4. Preferably, the current sink driver comprises a device, e.g. aMOSFET, provided on the current sink path, wherein the output of theamplifier provides the sink path voltage to control the current flowthrough the device.5. Additionally or alternatively, the current sink device comprises aplurality of output stages as part of the current sink path, preferablytwo, wherein the output stages are selectable based on a target sinkcurrent.6. Preferably, the plurality of output stages are configured to allowfor different levels of sink current to flow through the current sinkpath, as defined by the selected output stages.7. Preferably, the current sink device comprises a plurality of inputstages or configurations as part of the reference current path,preferably two, wherein the input stages or configurations areselectable based on the selected output stage.8. Additionally or alternatively, the current sink driver comprises afeedback resistor and a separate sense resistor provided in the currentsink path, such that a current generation operation is separated from acurrent sensing operation.9. Preferably, a sensing node is defined between the feedback resistorand the sense resistor, wherein the feedback resistor is connectedbetween the feedback-connected second amplifier input and the sensingnode, and the sense resistor is connected between the sensing node andground, and wherein the sensing node provides a voltage output formonitoring of the current sink driver.10. In one aspect, the reference current in the current reference pathmay be provided by a variable current source, e.g. via an IDAC.11. In an additional or alternative aspect, the feedback resistor may beprovided as a variable resistance, e.g. an RDAC.12. There is provided a host device comprising a current sink driver asdescribed above.

1. Current control circuitry for controlling a current through a load,the current control circuitry comprising: amplifier circuitry; referencevoltage generator circuitry configured to supply a fixed referencevoltage to a first input of the amplifier circuitry; an output stagecomprising: a control terminal coupled to an output of the amplifiercircuitry; a current input terminal configured to be coupled to theload; and a current output terminal; a variable resistance coupled tothe current output terminal of the output stage; and a feedback pathbetween the current output terminal of the output stage and a secondterminal of the amplifier circuitry for providing a feedback voltage toa second input of the amplifier circuitry.
 2. Current control circuitryaccording to claim 1, wherein the reference voltage generator circuitrycomprises: a current source configured to generate a fixed current; anda resistance.
 3. Current control circuitry according to claim 1, whereinthe output stage comprises a MOSFET device.
 4. Current control circuitryaccording to claim 1, wherein a resistance value of the variableresistance is digitally controllable.
 5. Current control circuitryaccording to claim 1, wherein the variable resistance comprises aresistive digital to analogue converter (DAC), wherein a resistance ofthe resistive DAC is based on a digital code input to the resistive DAC.6. Current control circuitry according to claim 5, wherein the resistiveDAC is configured to receive a clock signal, and wherein a change in aresistance of the resistive DAC in response to a change in the digitalcode input to the resistive DAC is synchronised to the clock signal. 7.Current control circuitry according to claim 5, further comprising aswitch, wherein the switch is operable to couple the first and secondinputs of the amplifier circuitry during a change from one digital codeinput to the resistive DAC to another digital code input to theresistive DAC.
 8. Current control circuitry according to claim 1,further comprising a current sense resistor coupled in series betweenthe variable resistance and ground, wherein the current sense signal isconfigured to generate a signal indicative of the current through theload.
 9. Current control circuitry according to claim 8, furthercomprising processing circuitry configured to adjust the operation ofthe current control circuitry based on the signal indicative of thecurrent through the load.
 10. Current control circuitry according toclaim 9, wherein the processing circuitry is configured to reduce asupply voltage or power down the current control circuitry in responseto determining that the current though the load exceeds a predeterminedthreshold.
 11. Current control circuitry according to claim 9, whereinthe processing circuitry is configured to adjust an operating parameterof the current control circuitry to reduce an error between a measuredcurrent through the load and a predefined current value.
 12. Currentcontrol circuitry according to claim 11, wherein the processingcircuitry is operative to adjust one or more of: a gain of the amplifiercircuitry; a reference current that is used to generate the referencevoltage; a resistance value of a resistance that is used to generate thereference voltage a supply voltage to a portion of the current controlcircuitry; and a resistance value of the variable resistance. 13.Current control circuitry according to claim 8, wherein the currentsense resistor is of a different type than the variable resistance. 14.Current control circuitry according to claim 13, wherein the currentsense resistor is a TaN (tantalum nitride) resistor.
 15. Current controlcircuitry according to claim 1, wherein the current control circuitry isconfigured to receive a first power supply voltage for powering thereference voltage generator circuitry and a second power supply voltagefor supplying current to the load.
 16. Current control circuitryaccording to claim 1, wherein the load is a light emitting diode (LED).17. Current control circuitry for controlling a current through a load,the current control circuitry comprising: amplifier circuitry; referencevoltage generator circuitry configured to supply a reference voltage toa first input of the amplifier circuitry; an output stage comprising: acontrol terminal coupled to an output of the amplifier circuitry; acurrent input terminal configured to be coupled to the load; and acurrent output terminal; a feedback path between the current outputterminal of the output stage and a second terminal of the amplifiercircuitry for providing a feedback voltage to a second input of theamplifier circuitry; and a current sense resistor coupled to ground in acurrent path for the load current.
 18. An integrated circuit comprisingcurrent control circuitry according to claim
 1. 19. An electronic devicecomprising an integrated circuit according to claim
 18. 20. Anelectronic device according to claim 19, wherein the device comprises amobile telephone, a tablet or laptop computer, a wearable device, agaming device, a virtual reality or augmented reality device.